USPTO Patents Application 09864891

WORLD INTELLECTUAL PROPERTY ORGANIZATION
Internationa) Bureau

PCT

INTERNATIONAL APPLICATION PUBLISHED UNDER THE PATENT COOPERATION TREATY (PCT)

(51) International Patent Classification 6 :
A61B5AX)

Al

(11) Internationa) Publication Number:

WO 96/26670

(43) International Publication Date: 6 September 1996 (06.09.96)

(21) International Application Number: PCT/US96700975

(22) International Filing Date: 24 January 1996 (24.01.96)

(30) Priority Data:

08/394,870

27 February 1995 (27.02.95) US

(71) Applicant: MEDTRONIC, INC [US/USJ; 7000 Central Av-

enue Northeast, Minneapolis, MN 55432 (US).

(72) Inventors: HALPERIN, Louis, E.; 1524 Goodrich Avenue,

SL Paul MN 55105 (US), MOESEL, Keith, A; 896
Ashland Avenue, St Paul, MN 55104 (US). UFFORD,
Keith, A^ 11337 Lekoday Lane, Chisago City, MN 55013
(US). SVENSK, James, 948 119th Lane Northwest,
Coon Rapids, MN 55448 (US). PATRICK, Timothy; 117
11th Avenue North, St Paul, MN 55075 (US). HASSLER.
Beth, Anne, H.; 1512 Park Street #8, White Bear Lake, MN
55110 (US). VARRICHIO, Anthony, J.; 1 Hemlock Lane,
Flanders, NJ 07836 (US).

(74) Agents: ATLASS, Michael, B. et al.; Medtronic, Inc., 7000
Central Avenue Northeast, MS 301, Minneapolis, MN 55432
(US).

(81) Designated States: AU, CA, JP, European patent (AT, BE,
CH, DE, DK. ES, FR. GB. GR, IE, IT. LU, MC. NL. FT,
SE).

Published

With international search report.

Before the expiration of the time limit for amending the
claims and to be republished in the event of the receipt of
amendments.

(54) Title: IMPLANTABLE CAPACTTIVE ABSOLUTE PRESSURE AND TEMPERATURE SENSOR

no

(57) Abstract

An endocardia] lead ( 12) for implantation in a right heart chamber (RV) for responding to blood pressure and temperature and providing
modulated pressure and temperature related signals to an implanted or external hemodynamic monitor (100) and/or cardiac pacemaker or
paamakei/cardiovertei/defjbru^tor. The lead (12) has a sensor module (20) formed in its distal end and is coupled to a monitor (100) mat
powers a sensor circuit in the sensor module. The sensor module is formed with a pickofT capacitor (Cp) mat changes capacitance with
pressure changes and a reference capacitor (Cp) that is relatively insensitive to pressure changes. The sensor circuit provides charge current
that changes with temperature variation at the implant site, alternately charges and discharges the two capacitors, and provides tuning
pulses having distinguishable parameters at the end of each charge cycle that are transmitted to a demodulator (150). The demodulator
(150) detects and synchronizes die timing pulses and derives pressure and temperature analog voltage values representative of the intervals
between the timing pulses which are digitized and stored in the monitor. The monitor may also be coupled with other sensors (106, 152)
for measuring and storing related patient body parameters, e.g. blood gas, ECG, and patient activity.

BEST AVAILABLE COPY

FOR THE PURPOSES OF INFORMATION ONLY

Codes used to identify States party to the PCT on the front pages of pamphlets publishing international
applications under the PCT.

AM

Armenia

GB

United Ktajdom

MW

Malawi

AT

Austria

GB

MX

Mexico

AU

Australia

GN

Guinea

NX

Niger

BB

Nnm

CR

Greece

NL

Netherlands

BE

Belgium

HU

Hungary

NO

Norway

BF

Burkina Paso

IE

Ireland

HZ

BG

Bulgaria

IT

Italy

PL

Poland

BJ

JP

Japan

FT

Portugal

BR

Brazil

KE

Kenya

RO

Romania

BY

Be tuts

KG

Kyrgystan

RU

Russian Federation

CA

Cauda

KP

DfiiMxiaiic People's Republic

SD

Sudan

CF

Centnl African Republic

SB

Sweden

CG

Cougp

KR

Republic of Korea

SG

CH

Switzerland

KZ

SI

Slovenia

a

CfitS tf*lVOCfC

U

SK

Slovakia

CM

LK

Sri Lanka

SN

Senegal

CN

ChMM

LR

Liberia

SZ

Swaziland

CS

LT

TD

Chad

CZ

Cash Republic

US

Luxemboorg

TG

Togo

DE

GcilUAUJf

LV

Latvia

TJ

Ta£kittao

DC

MC

TT

Trinidad and Tobago

KB

Estonia

MD

Repubfic of Moldova

UA

Ukraine

BS

Spain

MG

UO

Uganda

FI

Finland

ML

MaU

US

United States of Amc

FR

Prance

MN

Mongolia

uz

Uzbekistan

CA

Gaboo

MR

VN

VktNam

WO 96/26670

1

PCT/US96/00975

IMPLANTABLE CAPACITIVE ABSOLUTE

PRESSURE AND TFMPFHATIJRE SENSOR

5 BACKGROUND OF THF INVENTION

Field of the Invention . The present invention relates to a body implantable pressure
sensor, particularly attached to an endocardial lead for implantation in a right heart
chamber, for responding to blood and atmospheric pressure and blood temperature and
providing modulated pressure and temperature related signals to an implanted or
10 external hemodynamic monitor and/or cardiac pacemaker or
pacemaker/cardioverter/defibrillator.

Description of the Background Art. Efforts have been underway for many years to
develop implantable pressure transducers and sensors for temporary or chronic use in
a body organ or vessel. Many different designs and operating systems have been
15 proposed and placed into temporary or chronic use with patients. Indwelling pressure
sensors for temporary use of a few days or weeks are available, and many designs of
chronically or permanently implantable pressure sensors have been placed in clinical
use.

Piezoelectric crystal or piezoresistive pressure transducers mounted at or near
20 the distal tips of pacing leads, for pacing applications, or catheters for monitoring
applications, are described in U.S. Patent Nos. 4,407,296, 4,432,372, 4,485,813,
4,858,615, 4,967,755, and 5,324,326, and PCT Publication No. WO 94/13200, for
example. The desirable characteristics and applications for patient use of such lead or
catheter bearing, indwelling pressure sensors are described in these and other patents
25 and the literature in the field. Generally, the piezoelectric or piezoresistive

transducers have to be sealed hermetically from blood. Certain of these patents, e.g.
the * 296 patent, disclose sealing the piezoresistive bridge elements within an oil filled
chamber.

U.S. Patent No. 4,023,562 describes a piezoresistive bridge of four,
3 0 orthogonally disposed, semiconductor strain gauges formed interiorly on a single

crystal silicon diaphragm area of a silicon base. A protective silicon cover is bonded
to the base around the periphery of the diaphragm area to form a sealed, evacuated
chamber. Deflection of the diaphragm due to ambient pressure changes is detected by
the changes in resistance of the strain gauges.

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PCTAJS96/00975

Because the change in resistance is so small, a high current is required to
detect the voltage change due to the resistance change. The high current requirements
render the piezoresistive bridge unsuitable for long term use with an implanted power
source. High gain amplifiers that are subject to drift over time are also required to
amplify the resistance-related voltage change.

Other semiconductor sensors employ CMOS IC technology in the fabrication
of pressure responsive silicon diaphragm bearing capacitive plates that are spaced from
stationary plates. The change in capacitance due to pressure waves acting on the
diaphragm is measured, typically through a bridge circuit, as disclosed, for example,
in the article "A Design of Capacitive Pressure Transducer" by Ko et al., in IEEE
Proc Symp. Biosensors . 1984, p.32. Again, fabrication for long term implantation
and stability is complicated.

In addition, differential capacitive plate, fluid filled pressure transducers
employing thin metal or ceramic diaphragms have also been proposed for large scale
industrial process control applications as disclosed, for example, in the article "A
ceramic differential-pressure transducer N by Graeger et al., Philips Tech. Rev. .
43:4:86-93, Feb. 1987. The large scale of such pressure transducers does not lend
itself to miniaturization for chronic implantation.

Despite the considerable effort that has been expended in designing such
pressure sensors, a need exists for a body implantable, durable, long-lived and low
power pressure sensor for accurately sensing absolute pressure waves and related
parameters in the body over many years.

SUMMARY OF THE INVENTION

Accordingly, it is an object of the present invention to provide a practical and
durable indwelling absolute pressure sensor particularly for measuring blood pressure.

In accordance with the invention an implantable capacitive pressure sensor lead
for providing signals representative of the magnitude of body fluid pressure at a
selected site and ambient operating conditions, including body temperature, at the site
comprises: an elongated implantable lead body having proximal and distal end
sections and having first and second electrical conductors extending from the proximal
end section to the distal end section, the proximal end adapted to be coupled to a
biasing signal source, and the distal end section adapted to be implanted in a body

WO 96/26670

PCT/US96/00975

position for measuring a varying body fluid pressure; and a pressure sensor module
formed in the distal end section of the lead body and coupled to the first and second
electrical conductors, the pressure sensor module further comprising: a module
housing attached to the lead body adapted to be positioned in a body cavity and
5 enclosing a hermetically sealed chamber; pressure deformable pickoff capacitor

means for varying in capacitance in response to variations in fluid pressure having a
first plate formed of a pressure deformable diaphragm of the module housing and a
second plate spaced apart therefrom; reference capacitor means having a first plate
formed of a non-deformable portion of the module housing and a second plate spaced

10 apart therefrom; a substrate supported within the module housing; pressure and
temperature signal modulating circuit means supported on the substrate within the
hermetically sealed chamber and electrically coupled to the first and second electrical
conductors, the circuit means comprising means for receiving an externally applied
biasing signal and generating charging and discharging currents therefrom which vary

15 with ambient temperature, means for alternately charging and discharging the pickoff
and reference capacitor means, and means for generating pickoff and reference timing
pulses on one of the electrical conductors, the timing pulses separating the alternate
charging time intervals of the pickoff and reference capacitor means.

The pressure sensor lead is preferably constructed such that the first plate of

20 the pickoff capacitor means is formed of a pressure deformable, planar diaphragm
having a first predetermined surface area formed by the module housing of a
conductive material with an exterior planar surface and a parallel interior surface
within the hermetically sealed chamber, the planar diaphragm adapted to be deformed
in the first predetermined surface area by variations in fluid pressure outside the

2 5 module housing; and the first plate of the reference capacitor means is formed by a

second predetermined planar surface area of the interior surface spaced apart from and
co-planar with the first predetermined surface area and substantially non-deformable
by variations in fluid pressure outside the module housing. Plated standoffs are
employed to separate the first and second plates of the reference and pickoff capacitor

30 means.

The entire lead and module housing are electrically insulated from the body.
The insulation over the exterior surface of the diaphragm is effected by a thin,

WO 96/26670

PCT/US96/00975

strongly adhering material that prevents separation and the formation of voids which
could fill with liquid and change the response characteristics of the diaphragm to
pressure changes.

The construction of the reference capacitor employing an extension of the
5 common plate with the pickoff capacitor and the other plates on the same substrate,

with the plates separated by the same separator standoff platings simplifies fabrication.
The use of the temperature dependent charging current provides for the determination
of the temperature signal which can then be used to derive an absolute pressure signal
with the common mode temperature variations accounted for.
10 BRIEF DESCRIPTION OF THE DRAWING

These and other objects, advantages and features of the present invention will
be more readily understood from the following detailed description of the preferred
embodiments thereof, when considered in conjunction with the drawings, in which like
reference numerals indicate identical structures throughout the several views, and
15 wherein:

Figure 1 is block level diagram of an implantable, programmable blood
pressure monitor and lead system of the present invention;

Figure 2 is a cross-section assembly view of the distal end of a pressure
sensing lead employed in the system of Figure 1 ;
20 Figure 3 is a cross-section assembly view of the proximal end of a pressure

sensing lead employed in the system of Figure 1;

Figure 4 is a top subassembly drawing of the pressure sensing module
incorporated into the distal end of the pressure sensing lead of Figure 2;

Figure 5 is a side cross-section view of the internal components of the pressure
25 sensing module taken along line A- A of Figure 4;

Figure 6 is a partial cutaway perspective view of the attachment of the
pressure and temperature signal modulating circuit to the feedthrough and housing of
the pressure sensing module;

Figure 7 is an exploded perspective view of the components of the pressure
3 0 sensing module ;

Figure 8 is a side cross section view of a housing member and diaphragm taken
along line B-B of Figure 7;

WO 96/26670

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PCT/US96/00975

Figure 9 is a bottom view of the IC hybrid circuit substrate of Figure 7;
Figure 10 is a schematic diagram of the pressure sensing lead impedance
network and the pressure and temperature signal modulating circuit;

Figure 1 1 is a schematic diagram of the pressure and temperature signal

5 modulating circuit;

Figure 12 is a timing diagram of the pulse signals emitted by the circuits of
Figures 11 and 13; and

Figure 13 is a schematic diagram of the demodulator of the pressure monitor of

Figure 1.

10 DETAILED DE SCRIPTION OF THE PREFERRED EMBODIMENTS

The capacitive pressure sensing lead 12 of the present invention is designed to
chronically transduce blood pressure from the right ventricle of the heart over the
range of absolute pressures from 400-900 mm Hg, and within the frequency range of
0-100 Hz. The lead 12 is primarily employed with an implantable, battery powered

15 monitor 100 which employs a microprocessor based demodulation, data storage and
telemetry system for sampling and storing blood pressure data at programmed
intervals and telemetering out the accumulated data to an external
programmer/transceiver on receipt of a programmed-in command, in the manner of
current, conventional multi-programmable pacemaker technology. The lead 12 is

20 intended to be implanted transvenously into the right heart chambers in the same
manner as a conventional pacing lead, except that the distal end, including the
pressure sensor module, may be advanced out of the right ventricle into the pulmonary
artery to monitor blood pressure in that location. The monitor is intended to be
implanted subcutaneously in the same manner that pacemakers are implanted.

25 Monitor and Lead System Overview

Figure 1 is a simplified block diagram of the patient's heart 10 in relation to
the pressure sensing lead 12 and monitor 100. The lead 12 has first and second lead
conductors 14 and 16 extending from a proximal connector end 18 to the pressure
sensor module 20 disposed near the distal tine assembly 26. The pressure sensor
3 0 module 20 includes a variable pickoff capacitor and a fixed reference capacitor and
signal modulating circuit described below in reference to Figures 4-12 which
develops both blood pressure and temperature time-modulated intervals. The proximal

WO 96/26670 6 PCT/US96/00975

connector assembly is formed as a conventional bipolar, in-line pacing lead connector
and is coupled to the monitor connector (not shown) which is formed as a conventional
bipolar in-line pacemaker pulse generator connector block assembly. The tine
assembly 26 comprises soft pliant tines adapted to catch in heart tissue to stabilize the
5 lead in a manner well known in the pacing art. The detailed construction of the lead
12 is described below in conjunction with Figures 2 and 3.

The monitor 100 is divided generally into an input/output circuit 112 coupled
to a battery 108, an optional activity sensor 106, a telemetry antenna 134, the lead
conductors 14, 16, a crystal 110, and a microcomputer circuit 114. The input/output

10 circuit 112 includes the digital controller/timer circuit 132 and the associated

components including the crystal oscillator 138, power-on-reset (POR) circuit 148,
Vref/BIAS circuit 140, ADC/MUX circuit 142, RF transmitter/receiver circuit 136,
optional activity circuit 152 and pressure signal demodulator 150.

Crystal oscillator circuit 138 and crystal 110 provide the basic timing clock for

15 the digital controller/timer circuit 132. Vref/BIAS circuit 140 generates stable voltage
reference Vref and current levels from battery 108 for the circuits within the digital
controller/timer circuit 132, and the other identified circuits including microcomputer
circuit 114 and demodulator 150. Power-on-reset circuit 148 responds to initial
connection of the circuitry to the battery 108 for defining an initial operating condition

20 and also resets the operating condition in response to detection of a low battery voltage
condition. Analog-to-digital converter (ADC) and multiplexor circuit 142 digitizes
analog signals Vprs and Vtemp received by digital controller/timer circuit 132 from
demodulator 150 for storage by microcomputer circuit 114.

Data signals transmitted out through RF transmitter/receiver circuit 136 during

25 telemetry are multiplexed by ADC/MUX circuit 142. Voltage reference and bias

circuit 140, ADC/MUX circuit 142, POR circuit 148, crystal oscillator circuit 138 and
optional activity circuit 152 may correspond to any of those presently used in current
marketed, implantable cardiac pacemakers.

The digital controller/timer circuit 132 includes a set of timers and associated

30 logic circuits connected with the microcomputer circuit 1 14 through the data

communications bus 130. Microcomputer circuit 114 contains an on-board chip
including microprocessor 120, associated system clock 122, and on-board RAM and

WO 96/26670 7 PCTAJS96/00975

ROM chips 124 and 126, respectively. In addition, microcomputer circuit 114
includes an off-board circuit 118 including separate RAM/ROM chip 128 to provide
additional memory capacity. Microprocessor 120 is interrupt driven, operating in a
reduced power consumption mode normally, and awakened in response to defined
5 interrupt events, which may include the periodic timing out of data sampling intervals
for storage of monitored data, the transfer of triggering and data signals on the bus
130 and the receipt of programming signals. A real time clock and calendar function
may also be included to correlate stored data to time and date.

In a further variation, provision may be made for the patient to initiate storage

10 of the monitored data through an external programmer or a reed switch closure when
an unusual event or symptom is experienced. The monitored data may be related to an
event marker on later telemetry out and examination by the physician.

Microcomputer circuit 1 14 controls the operating functions of digital
controller/timer 132, specifying which timing intervals are employed, and controlling

15 the duration of the various timing intervals, via the bus 130. The specific current
operating modes and interval values are programmable. The programmed-in
parameter values and operating modes are received through the antenna 134,
demodulated in the RF transmitter/receiver circuit 136 and stored in RAM 124.
Data transmission to and from the external programmer (not shown) is

20 accomplished by means of the telemetry antenna 134 and the associated RF transmitter
and receiver 136, which serves both to demodulate received downlink telemetry and to
transmit uplink telemetry. For example, circuitry for demodulating and decoding
downlink telemetry may correspond to that disclosed in U.S. Patent No. 4,556,063
issued to Thompson et al. and U.S. Patent No. 4,257,423 issued to McDonald et al.,

25 while uplink telemetry functions may be provided according to U.S. Patent No.
5,127,404 issued to Wyborny et al. Uplink telemetry capabilities will typically
include the ability to transmit stored digital information as well as real time blood
pressure signals.

A number of power, timing and control signals described in greater detail
30 below are applied by the digital controller/timer circuit 132 to the demodulator 150 to
initiate and power the operation of the pressure sensor module 20 and selectively read
out the pressure and temperature signals Vprs and Vtemp. An active lead conductor

WO 96/26670 8 PCT7US96/00975

16 is attached through the connector block terminals to input and output terminals of
demodulator 150 which supplies a voltage VREG at the output terminal. A passive
lead conductor 14 is coupled through to the VDD supply terminal of the demodulator
150. The voltage signals Vprs and Vtemp developed from intervals between current
5 pulses received at the input terminal are provided by demodulator 150 to the digital
controller/timer circuit 132. The voltage signals Vprs and Vtemp are converted to
binary data in an ADC/MUX circuit 142 and stored in RAM/ROM unit 128 in a
manner well known in the art.

As depicted in Figure 1, the monitor 100 periodically stores digitized data

10 related to blood pressure and temperature at a nominal sampling frequency which may
be related to patient activity level, both optionally correlated to time and date and
patient initiated event markers. The monitor 100 may also optionally include a further
lead connector for connection with further lead for implantation in the right heart
having an exposed unipolar distal electrode from which an electrogram (EGM) may be

15 derived. The further lead may also have an oxygen sensor module in the distal

segment of the lead. Such a lead is shown in commonly assigned U.S. Patent No.
4,750,495 to Moore and Brum well, incorporated herein by reference. That
modification of the monitor 100 would also include an EGM sense amplifier (using the
monitor case as an indifferent electrode) and an oxygen sensor demodulator and is also

20 described in the above-incorporated x 495 patent.

In that optional configuration, the EGM signal may be employed to identify the
onset of a cardiac depolarization in each heart cycle and initiate either the monitoring
and storage operations or simply initiate the storage of the data derived by continuous
monitoring which would otherwise not be stored. In accordance with the preferred

25 embodiment of the invention, the monitored parameters including patient activity,

blood pressure and temperature, blood oxygen or other gas saturation level and EGM
are all continuously monitored.

The blood pressure signals are preferably digitized and stored at a sample
period of every 4.0 ms or 256 Hz sampling frequency. As shown below, this

3 0 frequency is about one-tenth of the operating frequency of the sensor module 20. The
blood temperature signals are preferably digitized and stored once during every sensed
EGM heart depolarization cycle. The digitized data values are stored on a FIFO basis

WO 96/26670 9 PCT/US96/00975

between periodic telemetry out of the stored data for permanent external storage.
Then, the data may be analyzed externally to identify the portions of the cardiac cycle
of interest and to perform other diagnostic analyses of the accumulated data.
Preferably, the external programmer can also program the monitor 100 to telemeter
5 out the digitized data values in real time at the same or different sampling frequencies.

The sampled and stored blood pressure data are absolute pressure values and
do not account for changes in barometric pressure affecting the ambient pressure load
on the pressure sensor module 20. Physicians typically measure blood pressure in
relation to atmospheric pressure. Thus, it may be necessary to separately record

10 atmospheric pressure data with separate measuring and recording equipment. At

present, a separate, portable pressure recording unit (not shown) worn externally by
the patient to record atmospheric pressure is contemplated to be used with the system
of the present invention. The atmospheric pressure and a time and date tag are
preferably recorded in the external unit at periodic, e.g. one minute, intervals. The

1 5 atmospheric pressure data is intended to be read out from the external unit when the
absolute pressure and optional other data stored in RAM/ROM unit 128 is telemetered
out and the data correlated by time and date and employed by the physician to derive
diagnoses of the patient's condition.
Pressure Sensor/Lead Construction

20 The pressure sensor capsule or module 20 is constructed with a titanium outer

housing having an integral, pressure defonnable, planar sensing membrane or
diaphragm formed in it as one plate of a variable or pickoff capacitor C P . The other
plate of the pickoff capacitor C P is fixed to one side of a hybrid circuit substrate
hermetically sealed within the outer housing. The capacitance of pickoff capacitor C P

2 5 varies as the diaphragm is deformed by pressure waves associated with heart beats in
the patient's heart 10 or elsewhere in the vascular system. A reference capacitor C R is
also formed with fixed spacing between planar capacitor plates formed on the same
side of the hybrid circuit substrate and on the outer housing to provide a reference
capacitance value. The pressure (and temperature) sensor circuitry within the module

30 20 employs the voltages VDD and VREG supplied by the demodulator 150 to

alternately charge and discharge the capacitor pair with charge and discharge currents
that vary with temperature and to provide instantaneous absolute pressure and

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PCT/US96/00975

temperature modulated charge time intervals to the demodulator 150 in a manner
described below.

Figures 2 and 3 are cross-section views of the distal and proximal end sections
of the lead 12 of Figure 1 . The pressure sensor module 20 is located just proximal to
the distal tip tine assembly 26 and is mechanically and electrically connected to the
coaxial, outer and inner, coiled wire lead conductors 14 and 16. The passive and
active, coiled wire lead conductors 14 and 16 are separated by an inner insulating
sleeve 22 and encased by an outer insulating sleeve 46 extending between in-line
connector assembly 30 and the pressure sensor module 20. A stylet receiving lumen
is formed within the inner coiled wire lead conductor 16 and extends to the connection
with the sensor module 20.

The in-line connector assembly 30 includes an inner connector pin 36 having a
stylet receiving, pin lumen 38 and is attached to the proximal end of the inner coiled
wire conductor 16 to align the pin lumen 38 with the stylet receiving lumen of the
inner coiled wire conductor 16. An insulating sleeve 40 extends distally over the
inner connector pin 36 and separates it from a connector ring 42. Connector ring 42
is electrically and mechanically attached to the proximal end of the outer coiled wire
conductor 14. An exterior insulating connector sleeve 24 extends distally from the
connector ring 42 and over the proximal end of the outer sleeve 46.

The distal ends of the outer and inner coiled wire conductors 14 and 16 are
attached to the proximal end of the pressure sensor module 20 to provide the VDD
and the input/output connections to the on-board pressure sensor hybrid circuit
described below. A further coiled wire segment 32 extends between the tine assembly
26 and the distal end of the pressure sensor module 20 and is covered by a further
insulating sleeve 34. The tine assembly 26 surrounds and electrically insulates inner
metal core 28 that is mechanically attached to the distal end of the coiled wire segment
32.

The materials used for these components of the pressure sensing lead 12 and
the construction and attachments depicted in Figures 2 and 3 are conventional in the
art of bipolar, coaxial pacing lead technology having an in-line connector. Such lead
technology is incorporated in the fabrication of the Medtronic 9 bipolar pacing lead
Model 4004M. The specific materials, design and construction details are not

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PCT/US96/00975

important to the understanding and practice of the present invention. In this particular
illustrated embodiment, the pressure sensing lead 12 is not also employed as a pacing
lead, but the pressure sensing module 20 could be incorporated into a pacing lead,
particularly for use in control of rate responsive pacemakers and pacemaker-
cardioverter-defibrillators .

Turning to the construction of the pressure sensing capsule or module 20,
reference is first made to the assembly drawings, including the enlarged top and side
cross-section views of Figures 4 and 5, the partial section view of Figure 6 and the
exploded view of Figure 7. The pressure sensing module 20 is formed with a first and
second titanium outer housing half members 50 and 52 which when joined together as
assembled titanium housing 55 surround a ceramic hybrid circuit substrate 60
supporting the sensing and reference capacitors and pressure signal modulating circuit.
The pressure signal modulating circuit (described in detail below with reference to
Figures 10 and 1 1) includes a resistor 62 and IC chip 64 mounted to one surface of the
substrate 60 and attached to electrical terminal pads and board feedthroughs to the
other surface thereof. The substrate 60 is supported in a manner described below in a
fixed relation with respect to housing member 52 by the proximal and distal silicone
rubber cushions 70 and 72 and the parallel side walls 47 and 49 (shown in Figures 7
and 8). The proximal silicone rubber cushion 70 also bears against the titanium
adaptor ring 74 that receives the feedthrough 76. The feedthrough 76 includes a
ceramic insulator 77 between the feedthrough ferrule 79 and feedthrough wire 80 to
electrically isolate feedthrough wire 80 that is electrically connected to a pad of the
substrate 60. The distal silicone rubber cushion 72 bears against the nose element 78
which is electrically connected to a further pad of the substrate 60.

Internal electrical connections between the sensor IC chip 64 and the substrate
are made via aluminum wire-bonds as shown in Figure 6. Connections between the
hybrid traces and both the feedthrough pin 80 and the nose element extension pin 75
are also made via gold wire-bonds. Conventional wire-bonding is used with each
trace, while the connections to the pins 75 and 80 are made using conductive silver
epoxy. The specific electrical connections are described below in conjunction with the
electrical schematic diagram of the sensor module electronic circuit in Figure 1 1 .

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PCT/US96/00975

After the mechanical and electrical components of the pressure sensing module
20 are assembled together, the titanium housing half members 50 and 52 and the nose
element and adaptor ring 74 are laser welded together as hermetically sealed,
assembled titanium housing 55. Then, the module 20 is attached to the components of
5 the lead 12 to provide the electrical and mechanical connections with the outer and
inner, passive and active, coiled wire lead conductors 14 and 16 as described below.

As shown in Figure 2, the module 20 is electrically and mechanically attached
to the outer and inner coiled wire conductors 14 and 16 at the proximal end thereof
through an intermediate transition assembly similar to a feedthrough and including an
10 insulating body 56 separating an inner, conductive transition pin 58 from distal and
proximal outer conductive transition sleeves 57 and 59. Sleeves 57 and 59 are laser
welded together for electrical and mechanical connection.

The distal transition sleeve 57 is welded to the ferrule 79 and the distal end of
the transition pin 58 is staked to the feedthrough pin 80. The distal end of the inner
1 5 transition pin 58 is hollow and extends out of the insulating body 56 to receive the

proximal end of the feedthrough pin 80. Staking is accomplished through access ports
in the molded insulating body 56, and then the access ports are filled with silicone
adhesive. In this fashion, the inner transition pin 58 is electrically coupled to the
feedthrough pin 80, and the outer transition sleeves 57 and 59 are electrically

2 0 connected to the assembled titanium housing 55.

The proximal end of the inner transition pin 58 is slipped into the distal lumen
of the inner coiled wire conductor 16. The distal end of the inner coiled wire
conductor 16 is crimped to the proximal end of the inner transition pin 58 by force
applied to a crimp sleeve 66 slipped over the distal segment of the coiled wire
25 conductor 16. The distal end of inner insulating sleeve 22 is extended over the crimp
sleeve 66 and adhered to the insulating body 56 to insulate the entire inner conductive
pathway. The outer coiled wire conductor 14 is attached electrically and mechanically
by crimping it between the outer transition sleeve 59 and an inner crimp core sleeve
68 slipped between the distal lumen of the outer coiled wire conductor 14 and the

3 0 inner insulating sleeve 22. Silicone adhesive may also be used during this assembly.

When the electrical and mechanical connections are made, the active coiled wire
conductor 16 is electrically connected to a pad or trace of the substrate 60, and the

WO 96/26670 13 PCT/DS96/00975

passive coiled wire conductor 14 is electrically attached through the housing half
members 50 and 52 to a further substrate pad or trace as described below.

The distal end of the pressure sensing module 20 is attached to the distal lead
assembly including further outer sleeve 34 and coiled wire conductor 32 described
5 above. At the distal end of the pressure sensing module 20, a crimp pin 81 is inserted
into the lumen of the further coiled wire conductor. The crimp pin 81 and the further
coiled wire conductor 32 are inserted into the tubular nose element 78 which is then
crimped to the coiled wire conductor 32 and crimp pin 81. The further outer sleeve
34 extends over the crimp region and the length of the further coiled wire conductor

10 32. The distal end of the farther coiled wire conductor 32 is attached by a similar
crimp to the inner tip core member 28 using a further crimp pin 27.

Returning to Figures 4 and 5, and in reference to Figure 8, thin titanium
diaphragm 54 is machined into the titanium outer housing half member 50. The flat
inner surface of diaphragm 54 and a peripheral continuation of that surface form plates

15 of a pair of planar capacitors, the other plates of which are deposited onto the adjacent
surface 61 of the ceramic hybrid substrate 60 as shown in Figure 9. An external
pressure change results in displacement of the diaphragm 54 and subsequent change in
capacitance between the diaphragm 54 and one of the deposited substrate plates. This
change in capacitance of the pickoff capacitor C P with change in pressure is

20 approximately linear over the pressure range of interest, and is used as a measure of
the pressure outside the sensor module 20. The external pressure change has little
effect on the second, reference capacitor C R .

To electrically isolate diaphragm 54 from the patient's body, materials must be
used that do not significantly absorb body fluids and swell, which in turn causes

2 5 diaphragm 54 deflection and changes the capacitance of the pickoff capacitor C P . The
material must be uniformly thin and repeatable during manufacture so as to avoid
affecting sensitivity of the pickoff capacitor C P . Also, the material must adhere very
well to the diaphragm 54 so that bubbles, gaps or other separations do not occur over
time. Such separations could cause hysteresis or lag of the sensed capacitance change.

30 Returning to Figure 2, the outer sleeve 46 and further sleeve 34 are formed of

conventional urethane tubes employed in fabricating pacing leads. For adherence to
outer sleeve 46 and further sleeve 34, a thin urethane sensor jacket or covering 82 is

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employed that extends over the full length of the sensor module 20 and is adhered at
its ends to the outer insulating sleeve 46 and the further outer insulating sleeve 34 ,
e.g. as by urethane based adhesives. The urethane covering 82 is employed to cover
the majority of the sensor module 20 but the material does not always adhere well to
the metal surfaces thereof, even when a primer is employed. The loss of adherence
over the diaphragm 54 can lead to accumulation of fluids and affect the response time
to changes in blood pressure. Therefore, it is necessary to substitute a better
adhering, body compatible, insulating coating over the diaphragm 54.

In order to do so, a cut-out portion of the sensor covering 82 is made following
the periphery 53 in order to expose the diaphragm or diaphragm 54. A thin, uniform
thickness coating 45 of silicone adhesive is applied over the exposed diaphragm 54
that adheres thereto without any fluid swelling or separation occurring over time. The
silicone adhesive does not adhere well to the edges of the cut-out section of the
urethane covering 82, but can be injected between the edges and the half member 50
to fill up any remaining edge gap.

The resulting composite covering 82 and insulating layer electrically insulates
the titanium outer housing half members 50 and 52 that are electrically connected to
VDD. The combined housing is formed by welding the half members 50 and 52
together and to the adaptor ring 74 and nose element 78. When assembled, the sensor
capsule or module 20 is preferably about 0.140 inches in diameter, including the
polyurethane insulation covering 82, and is approximately 0.49 inches long.

The cylindrical housing half members 50 and 52 are machined in the two
pieces using wire electric discharge machining (EDM) methods. In the first housing
half member 50, the thin diaphragm 54 is approximately 0.0013 inches thick at T in
Figure 8 and is produced through precision EDM of the interior and exterior surfaces
of the titanium stock. The inner surface 51 of the half member 50 extends as a
continuous planar surface beyond the perimeter 53 of the diaphragm 54 to provide one
plate of the reference capacitor C R in that region.

Turning to Figure 9, the ceramic sensor hybrid circuit substrate 60 consists of
a 90% alumina board, on the back side 61 of which are deposited an inner,
rectangular capacitor plate 84 coupled to a plated substrate feedthrough 98, an outer,
ring shaped capacitor plate 86 coupled to a plated substrate feedthrough 96, and three

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plated standoffs 88 t 90, 92. The inner capacitor plate 84 is dimensioned to generally
conform to the shape of the diaphragm 54 and fall within the perimeter 53. The
perimeter or ring-shaped capacitor plate 86 is dimensioned to fall outside or just inside
or to straddle the perimeter 53. The inner surface 51 of half member 50 provides a
reference surface for locating the capacitor plates 84 and 86 relative to the diaphragm
54.

When assembled, the plates 84 and 86 are spaced from the inner surface 51 of
the housing half member 50 by the difference in thicknesses of the standoffs 88 - 92
and the plates 84 and 86 to form the pickoff capacitor C P and reference capacitor C R .
The pressure sensing pickoff capacitor C P employing central capacitor plate 84 varies
in capacitance with pressure induced displacement of the diaphragm 54 and the
silicone adhesive layer applied thereto. The reference capacitor C Rt employing the
perimeter reference capacitor plate 86 located in the region where diaphragm 54
deflection is negligible within the operating pressure range, varies in capacitance with
common mode changes in sensor voltages, thermal expansion effects, and changes in
the hermetically sealed capacitor dielectric constant.

The two capacitor plates 84 and 86 are electrically connected to the front side
of the substrate 60, on which the sensor electronic circuit included in the IC chip 64
and the resistor 62 are mounted. The common capacitor plate surface 51 is coupled to
VDD. The sensor electronic circuit alternately charges and discharges the pickoff and
reference capacitors C P and C R through a constant current source which varies with
temperature change inside the sensor module 20. The temperature-related changes in
the charging current affects the charge times for both the pickoff and reference
capacitors C P and C R equally. However, temperature induced changes in internal
pressure within the sensor module 20 (and external pressure changes) only affect the
pickoff capacitor C P plate spacing, which causes an increase or decrease in the
capacitance and subsequent increase or decrease in the time to charge the pickoff
capacitor C P to a set voltage level.

Blood pressure changes cause an increase or decrease of the pickoff capacitor
C P plate spacing, which causes a decrease or increase, respectively, in the capacitance
and subsequent decrease or increase, respectively, in the time to charge the pickoff
capacitor C P to a set voltage level, assuming an unchanged blood temperature and

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constant charging current. Since no significant gap change between common plate
surface 51 and the perimeter capacitor plate 86 due to pressure change occurs at the
reference capacitor C R , there is little pressure induced reference capacitance change.
The ratio of the charging time of the pickoff capacitor C P to the sum charging time of
the reference and pickoff capacitors C R and C P provides a stable indication of pressure
induced changes and cancels out common mode capacitance changes, resulting in an
absolute pressure signal. The common mode capacitance change, principally
temperature related, can be derived from the capacitance of the reference capacitor
C R . The signals Vprs and Vtemp can also be time correlated to EGM, activity signals
and blood gas signals and all stored in memory for later telemetry out.

The substrate surface 61 platings shown in Figure 9 are specially designed to
provide precise control of the pickoff and reference capacitor gaps without the need
for an excessive number of close-tolerance components. By specifying a single tight
tolerance between the top surfaces of the standoff platings and the top surfaces of the
capacitor platings, the spacing between the reference and pickoff capacitor plates and
the planar surface 51 of the sensor diaphragm 54 can be very accurately controlled.
Because the inner surface 51 of the diaphragm 54 extends beyond the perimeter 53 of
the diaphragm 54 to the region where the standoffs 88 - 92 make contact, the
difference between the height of the standoff pads and the height of the capacitor
plates 84 and 86 will define the gap between the capacitor plates 84, 86 and the inner
surface 51.

The hybrid standoffs 88 - 92 are pressed into contact against the inner surface
51 by the compressible molded silicone rubber cushions 70, 72 when the components
are assembled. The assembly creates an interference fit exerting pressure between the
surface 51 and the standoffs 88-92. Lateral constraint of the substrate 60 is provided
by the fit of the hybrid circuit substrate 60 in the housing half member 50 between the
lateral side walls 47 and 49 in one axis, and by the silicone rubber cushions 70, 72
along the other axis. In a variation, the rubber cushions 70, 72 may be eliminated in
favor of simply adhering the side edges of the substrate 60 to the side walls 47 and 49
and/or the end edges to the inner surface 51 of the half member 50. Or the adhesive
could be used with the rubber cushions 70, 72. Furthermore, in each assembly
variation, the standoffs 88 - 92 could be bonded to the inner surface 51. The result is

WO 96/26670 17 PCT/US96/00975

an accurate and permanent location of the substrate 60 within the cavity of the sensor
module 20 with no residual stress in the critical parts which might cause drift of the
sensor signal over time.

This approach to spacing the pickoff and reference capacitor C P and C R plates
5 has two major advantages. First, only one set of features, that is the plating heights or
thicknesses, need to be in close tolerance, and those features are produced through a
process which is extremely accurate. For, example, the standoffs 88, 90, 92 can be
precisely plated to a thickness of 0.0011, and the capacitor plates 84, 86 can be plated
to 0.005 inches. A gap of 0.0006 inches with a tolerance of 0.0001 inches can
10 thereby be attained between the capacitor plates 84, 86 and the diaphragm inner

surface 51. The second advantage is the near absence of signal modulation by thermal
expansion effects. Thermal change in dimension of the structure which establishes the
gap between the plates 84, 86 and the sensor diaphragm inner surface 51 is per the
relation D 1 = a DT 1, where D I is the change in gap, a is the thermal expansion
15 coefficient of the material creating the gap, DT is the variation in temperature, and 1
the length of the structure.

In the example provided, the gaps are only 0.0006" thick, so change in gap
over an expected variation in temperature in vivo of 1 0 C, assuming coefficient of
thermal expansion for the standoff material of around 13 x 10" 6 /°C, would result in a
20 gap change of 7.8 nano-inches. This thermal change is about sixty times less than the
gap change for 1 mm Hg pressure change, and much less than be detected using state
of the art low-current methods.

There is one significant thermal effect. When the pressure sensor module 20 is
sealed using a laser weld process, a volume of gas (mostly Argon and Nitrogen) at or

2 5 near atmospheric temperature and pressure is trapped inside the cavity of the sensor

module 20. The difference between the gas pressure inside the cavity and the outside
pressure influences the gap of the pickoff capacitor C P . At the instant the sensor is
sealed, there is zero pressure differential and consequently no deflection of the
pressure diaphragm 54 from its neutral position. But the gas inside the sensor must

3 0 comply with the classical gas law PV = nRT. Assuming then that the volume inside

the sensor is constant, and that the mass quantity and gas constant (n and R,
respectively) are constant (since no gas enters or leaves the sensor cavity after

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sealing), the effect of temperature change can be described by the gas law formula as
P 2 = Pj (T 2 /T,).

In the human body, and particularly the venous blood stream in the ventricle,
the temperature may vary from the nominal 37 °C (±2 °C). The variation may be
5 between ±3 °C with fever and between -1 °C to +2 °C with exercise. Assuming that
the sensor were sealed at 300K and 760 mm Hg, the gas law formula implies that for
every 1 ° C change in temperature there is a corresponding change of over 2 mm Hg in
internal pressure. This will manifest itself as a decrease in the pressure value reported
by the sensor with increasing temperature, since the cavity pressure against which

10 external pressure is compared has increased. This is a significant error and needs to
be compensated for. In accordance with a further aspect of the present invention, the
charging time of the reference capacitor C R , which will vary as a function of
temperature due to variation of band-gap regulator current of approximately 1 % / °C,
is monitored. The change in charging time Ttemp of the reference capacitor C R is

15 stored in the monitor 100, and used to correct for changing temperature effects.

As previously mentioned, the feature which physically responds to pressure to
produce a change in pickoff capacitance is the thin diaphragm 54 in housing half
member 50 created via a wire EDM process. The deflection y measured at the center
of the diaphragm 54 is governed by the equation:

20 ymax = k, (wr 4 / Et 3 )

where w is the pressure applied to one side of the diaphragm 54 (or the pressure
difference), r is the width of the rectangular diaphragm 54, E is Young's Modulus for
the diaphragm material, t is the thickness of the diaphragm, and kj is a constant
determined by the length-to-width ratio of the diaphragm 54. In the present invention,

25 a ratio of 2:1 was used for the sensor diaphragm 54 dimensions, yielding kj = .0277.
If the ratio were reduced to 1.5:1, and width remained constant, k t would be reduced
to .024, with a corresponding 13% reduction in diaphragm displacement. This is not
a major impact on sensitivity, and shows that the length of the diaphragm 54 could
potentially be reduced without a major impact on sensitivity.

30 In a specific construction employing the 2: 1 ratio and a diaphragm thickness T

of 0.0013 inches, and a gap of 0.005 inches, a baseline capacitance of approximately 3
pF was realized for both the pickoff and reference capacitors, C P and C R counting a

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capacitance contribution of the sensor IC chip 64. Baseline capacitance is preferably
large in comparison to expected parasitic capacitances, especially those which would
tend to vary over time or in response to environments, but not so large as to demand
overly large charging currents. Also, the capacitances are preferably large enough to
5 keep the oscillation frequency of the pickoff circuit around 4-6 kHz without resorting
to extremely low charging currents, which would tend to decrease signal-to-noise
ratio. Preliminary prediction for change in capacitance in response to pressure change
is 0.5- 1.5fF/mmHg.

The preferred embodiment of the reference and pickoff capacitors described

10 above and depicted in the drawings, particularly Figure 9, positions the reference
capacitor plate 86 in a ring shape surrounding the pickoff capacitor plate 84 on
substrate surface 61. It will be understood that the reference capacitor plate 86 may
have a different shape and be positioned elsewhere on the substrate surface 61. For
example, both the reference capacitor plate 86 and the pickoff capacitor plate 84 may

15 be square or rectangular and positioned side by side on the substrate surface 61 .

Regardless of the configuration or position, the reference capacitor plate would be
located outside the perimeter 53 of the diaphragm 54 and spaced away from the inner
surface of the diaphragm 54 in the same fashion as described above. Moreover, in
any such configuration, the diaphragm 54 and the pickoff capacitor plate 84 may also

20 have a different shape, e.g. a more square shape than shown.
Pressure and Te mperature Signal Modulating Circuit

The pressure and temperature signal modulating sensor circuit 200 (including
the circuit within the IC chip 64, the associated resistor 62 mounted on the substrate
60 and the pickoff and reference capacitors C P and C R ) within pressure sensing module

25 20 is shown in Figures 10 and 11. Sensor circuit 200 translates the pressure and

temperature modulated pickoff and reference capacitor C P and C R values into charge
time-modulated intervals Tprs and Ttemp, respectively, between sensor current pulse
signals P R and P P , transmitted up the active lead conductor 16.

Figure 10 also depicts the equivalent circuit impedance of the pressure sensing

30 lead 12 within the dotted line block denoted 12. The lead conductors 14 and 16 can
exhibit a leakage resistance 202 as low as about 300kW and capacitance 204 of about
1 10 pf between them. Lead conductor 14 has a series resistances 206 and 208 totaling

WO 96/26670 20 PCT/US96/00975

about 25 W, and lead conductor 16 has a series resistances 210 and 212 totaling about
40W. The leakage resistance and capacitance may deviate over the time of chronic
implantation. The demodulator 150 includes lead load impedances and is calibrated at
implantation in a manner described below.
5 The passive lead conductor 14 applies VDD from demodulator 150 to the VDD

terminal of IC chip 64 and to the pickoff and reference capacitors C P and C R . The
active lead conductor 16 connects the terminal VREG of IC chip 64 to the terminals
CPOUT and CPIN of demodulator 150 through an equivalent resistor network
depicted in Figure 13.

1 0 The pressure and temperature signal modulating sensor circuit 200 is shown in

greater detail in Figure 1 1 and essentially operates as a bi-stable multivibrator
operating near the target frequency of 5 kHz in alternately charging plate 86 of
reference capacitor C R and plate 84 of the pickoff capacitor C P from VDD, which in
this case is 0 volts, through reference voltage VR and to a target voltage VT through a

15 current source of 1/3 I as shown in the two waveforms of Figure 12 labeled VC R and
VC P . The reference capacitor C R and the pickoff capacitor C P are alternately
discharged through a further current source of 2/3 1 coupled to VDD through the
reference voltage VR back to VDD or 0 volts as also shown in these two waveforms
of Figure 12. It should be noted that the wave forms of Figure 12 are not to scale and

20 are exaggerated to ease illustration of the signals generated in the sensor circuit 200
and the demodulator circuit 150.

The pickoff and reference capacitors C P and C R are both nominally 2.2 pF,
but approach 3.0 pF with stray capacitances. Due to the biasing convention
employed, the reference capacitor C R and the pickoff capacitor C P are considered to be

25 discharged when their plates 86 and 84, respectively, are both at VDD or 0 volts.
The common plate 51 is always at VDD or 0 volts. The reference and pickoff
capacitors C R and C P are considered to be charged (to some charge level) when the
plates 84 and 86 are at a voltage other than 0 volts. In this case, the charges are
negative charges between VDD and VREG or between 0 and -2.0 volts. Thus, the

3 0 convention employed dictates that reference and pickoff capacitors C R and C P are

"charged" toward -2.0 volts and "discharged" from a negative voltage toward 0 volts.

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The principle involved is also applicable to a VSS convention, where the charged
voltage levels would be positive rather than negative in polarity.

In practice, when demodulator 150 of Figure 13 is powered up, it supplies the
voltage VDD at 0 volts to lead conductor 14 and VREG at -2.0 volts to lead conductor
16 of the lead 12. The regulated voltages VDD and VREG supplied by the
demodulator 150 to the sensor 200 of Figure 11 are applied to a voltage dividing diode
network including diodes 214, 216, and 218 and current source 232 in a first branch,
diode 220, external resistor 62, and current source 234 in a second branch, and diode
222 and current source 236 in a third branch. Voltage VT is three diode forward
voltage drops lower than VDD through diodes 214, 216 and 218, or about -1.5 volts,
and voltage VR is two diode forward voltage drops lower than VDD through diodes
214 and 216 or about -1.0 volts.

Differential current amplifier 230 is coupled to the second and third branches
and its output is applied to current sources 232, 234 and 236 in each branch. The
current I is defined by the voltage difference between two diodes 220 and 222
operating at significantly different current densities, divided by the value of the chip
resistor 62. Changes in ambient temperature affect the diode resistances and are
reflected in the output signal from differential amplifier 230. Current sources 234,
236 are driven to correct any current imbalance, and current source 232 develops the
current I reflecting the temperature change within the sensor module 20.

The principle employed in the pressure and temperature signal modulating
sensor circuit 200 is a deliberate misuse of the band gap regulator concept, in that
rather than using the band gap method to create a current source insensitive to
temperature, the current source 232 varies a known amount, about 1 % / °C, with
variation in temperature. This allows the variation in the reference capacitor C R
charge-modulated time Ttemp to be used as a thermometer, in the interest of
correcting for sensor internal pressure change with temperature and subsequent
absolute pressure error affecting the gap, and hence the capacitance, of the pickoff
capacitor C P . Since the gap of the reference capacitor C R cannot change significantly
with pressure or temperature, the primary change in Ttemp can only occur due to
temperature induced change in current I generated by current source 232.

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The reference voltage VR and the target voltage VT are applied to the switched
terminals of schematically illustrated semiconductor switches 258 and 260. The
common terminals of semiconductor switches 258 and 260 are coupled to a positive
input of comparators 240 and 242, respectively. The negative terminals of
comparators 240 and 242 are coupled through the series charge resistors 244 and 246,
respectively, to the plates 84 and 86 of the pickoff capacitor C P and the reference
capacitor C R , respectively. The outputs of the comparators 240 and 242 are inverted
by inverters 248 and 250, respectively, and applied to inputs of the flip-flop 252. The
outputs of the flip-flop 252 are applied to control terminals of the schematically
illustrated semiconductor switches 254 and 256. Semiconductor switches 254 and 256
are bistable in behavior and alternately connect current source 272, providing 2/3 I,
and current source 274, providing 1/3 I, to the reference capacitor C R and the pickoff
capacitor C P depending on the state of flip-flop 252. When the current source 272 is
applied to one of the capacitors, the current source 274 is applied to the other
capacitor. The capacitor voltage on plate 84 or 86 is discharged through current
source 272 back to VDD or 0 volts while the capacitor voltage on plate 86 or 84,
respectively, is charged through current source 274 toward VT as shown in Figure 12.

The outputs of comparators 240 and 242 are also applied to control the states
of schematically illustrated semiconductor switches 258, 260, 262, 263 and 264.
Semiconductor switches 258 and 260 are monostable in behavior and switch states
from the depicted connection with target voltage VT to reference voltage VR each
time, and only so long as, a high state output signal is generated by the respective
comparators 240 and 242. The timing states of these switches 258 and 260 closed for
conducting VR or VT to respective comparators 240 and 242 are also shown in the
wave forms labeled 258 and 260 shown in Figure 12.

The outputs of comparators 240 and 242 are normally low when the capacitor
charge voltages VC P and VC R , respectively, applied to the positive terminals are
lower, in an absolute sense, than the voltages VT applied to the negative terminals.
The charging of the capacitor C P or C R coupled to the charge current source 274 to the
voltage VT or -1.5 volts causes the associated comparator 240 or 242 to go high.
When the comparator goes high, the flip-flop 252 changes state exchanging the closed
states of semiconductor switches 254 and 256, thereby causing the previously charging

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(or fully charged) capacitor to commence discharging and causing the previously
discharged other capacitor to commence charging.

The high output state of the associated comparator remains for a predetermined
capacitor discharge time period from VT to VR providing a one-shot type, high state
5 output. When the capacitor C P or C R voltage discharges to VR, the high state output
of the respective comparator 242 or 240 is extinguished, and semi-conductor switches
258 or 260 is switched back to apply VT to the respective negative terminal of the
comparator 240 or 242. However, the capacitor C P or C R continues to discharge until
the plate 84 or 86, respectively, is back at full discharge or 0 volts. Since the

10 discharge rate exceeds the charge rate, there is a period of time in each cycle that the
capacitor C P or C R remains at 0 volts while the other capacitor charges toward VT (as
shown in Figure 12). This ensures that each capacitor is fully discharged to 0 volts at
the start of its respective charge time interval.

As shown specifically in Figure 1 1 , the switches 258 and 260 are set to apply

15 the voltage VT to comparators 240 and 242, and the switches 262, 263 and 264 are all
open. The plate 84 of pickoff capacitor C P is connected with the 2/3 I current source
272 and is being discharged toward VDD, that is 0 volts, while the plate 86 of
reference capacitor C R is connected with the 1/3 I current source 274 and is being .
charged toward VT or -1.5 volts. Because of the arrangement of the switches, 258,

2 o 260, 262, 263, and 264, no pulses are being generated. It can be assumed that the
plate 84 of pickoff capacitor C P is being charged from VDD toward VR and that the
voltage on the plate 86 of reference capacitor C R is discharging from VR toward
VDD. When the output of comparator 242 does go high, the high state signal will
cause switch 260 to switch over from the then closed pole position (e.g. the pole

2 5 position schematically depicted in Figure 1 1) to the other open pole position and

remain there until the comparator 242 output goes low again when the capacitor
voltage falls back to VT. Similarly, when the output of comparator 240 goes high in
the following charge cycle, the high state signal causes switch 258 to switch over from
the then closed pole position (e.g. the pole position schematically depicted in Figure

3 0 1 1) to the other pole position and remain there until the comparator 240 goes low. In

this fashion, the reference voltage VR is alternately applied by switches 258 and 260,

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respectively, to the negative terminals of comparators 240 and 242 for the relatively
short VR to VT discharge times shown in Figure 12.

To summarize, when the charge voltage on the pickoff capacitor C P reaches
VT, the comparator 240 switches its output state high, in turn changing the state
switch 258 and closing switch 263. Delay circuit 270 is enabled to close switch 264
when switch 263 re-opens. Similarly, in the next cycle, when the voltage on reference
capacitor C R reaches VT, the comparator 242 switches its output state high, changing
the state of switch 260 and closing switch 262.

Normally open semiconductor switches 262 and 263 are also monostable in
behavior and are closed for the duration of the comparator high state, that is, the VT
to VR discharge time period. When closed, the timing current pulses P R and P P
separating (at their leading edges) the reference and pickoff charge-time modulated
intervals Ttemp and Tprs also shown in Figure 12 are generated.

The timing current signal pulse P P is controlled in width by the reference
capacitor C R capacitor discharge time from VT to VR as shown in the Sensor Current
line of Figure 12. The initial low amplitude step of two step timing current signal
pulse P R is also controlled in width by the reference capacitor C R capacitor discharge
time from VT to VR as shown in the wave form labeled Sensor Current in Figure 12.
The VT to VR discharge times, which govern the closed time periods of switches 258
and 260 and the widths of the low amplitude steps of the timing current pulses P R and
P P , are nominally 8-12 msec. The high amplitude step of two step timing current
signal pulse P R is controlled in width by delay circuit 270 of Figure 11.

The high state signal output of comparator 242 therefore closes normally open
switch 262 for the duration of the high state, i.e. the VT to VR discharge time of
reference capacitor C R . When switch 262 closes, the current source 266 providing 64
I is applied to the VREG terminal, resulting in the generation of the timing current
pulse P P depicted in Figure 12 appearing on conductor 16 as a sensor current.
Similarly, the high state signal output of comparator 240 also closes the normally open
switch 263 for the VT to VR discharge time of duration of pickoff capacitor C P and is
applied to the delay circuit 270. Delay circuit 270 effects the closure of switch 264 at
the end of the high state and then maintains closure of the switch 264 through a
delayed high state time period. When switches 263 and 264 are sequentially closed,

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the current source 266 providing 64 I is applied to the VREG terminal, and then the
current source 268 providing 208 I is applied to the VREG terminal. In this manner,
the stepped current timing pulse P R depicted in Figure 12 is generated.

The nominal pulse height of 8.0 mA for timing current pulse P P and for the
initial step of timing current pulse P R is effected by the 64 I current source 266 when
either switch 262 or 263 is closed. The nominal, pulse height of 24.0 mA (stepped up
from the initial 8.0 mA step) of pulse P R is effected by the 208 I current source when
switch 264 is closed after switch 263 reopens. Between pulses, a baseline supply
current of 1 .5 mA is present at VREG and on lead conductor 16 to which the current
pulse heights or sensor current amplitudes are referenced.

The 8.0 mA leading step of pressure-related timing current pulse P P matches
the slew rate of the 8.0 mA peak of temperature-related, reference timing current
pulse P Rt which reduces errors that would otherwise be associated with detection of
different amplitude pulses having differing slew rates. The rise time of both of the
pulses appears to be the same to the current sensor 154 in the demodulator 150. The
start of each pulse can therefore be accurately detected and employed as the start and
end times for the intervening charge time intervals Tprs and Ttemp. The differing
peak amplitudes of the two pulses are readily distinguishable to determine the order of
the intervals.

Thus, Figure 12 illustrates the waveforms at the switches 258, 260, 262, 263
and 264 in relation to the charge and discharge voltage waveforms of the reference
and pickoff capacitor C R and C P as well as the timing current pulses P P and P R
generated at the terminal VREG marking the starts of the respective capacitor charging
intervals Tprs and Ttemp.

At 37° temperature and a barometric pressure of 740 mm Hg, the capacitance
values of capacitors C P and C R are approximately equal. Therefore, both capacitors
C P and C R charge at an approximately equal rate. The intervals between timing signal
pulses P P and P R are approximately equal, reflecting a 50% duty cycle (calculated as
the ratio of Tprs to Ttemp + Tprs), and the nominal operating frequency from P P to
P P is 5 kHz.

After implantation, the temperature should vary somewhat from 37°. The
current I, which changes with temperature change, affects the charge times Ttemp and

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Tprs equally which changes the operating frequency. In addition, the blood pressure
change between systole and diastole alters the capacitance of the pickoff capacitor C P
which only affects the charge time Tprs. Thus, charge time Ttemp only changes with
temperature, and the combined result is a change in frequency and duty cycle
dependent on both temperature and pressure changes.

The schematically illustrated current sources and semiconductor switches may
be readily realized with conventional integrated circuit designs.

Demodulator Circuit

The demodulator 150 shown in Figure 13 supplies the voltages VDD and
VREG, at a baseline current drain from sensor IC chip 64 of about 1 .5 mA, to the
lead conductors 14 and 16 and receives the timing signal current pulses P P and P R
modulating the baseline current on conductor 16. The demodulator 150 converts the
charge time intervals Tprs and Ttemp separating the leading edges of the train of
current pulses of Figure 12 into voltage signals Vprs and Vtemp, respectively. The
voltage signals Vprs and Vtemp are supplied to the digital controller/timer circuit 132
and are converted by ADC/MUX circuit 142 into digital values representing absolute
pressure and temperature data, which are stored in the microcomputer circuit 114 in a
timed relationship with other monitored physiologic data.

As described above, the analog temperature signal Vtemp is derived from the
interval Ttemp between the leading edges of P R and P P in an integration process, and
the analog pressure signal Vprs is derived from the interval Tprs between the leading
edges of P P and P R in a duty cycle signal filtering and averaging process. In these
processes, the demodulator 150 creates the intermediate voltage square waves
NCAPSOUT, NRESET OUT, and DCAPS shown in Figure 12 from the current
pulse timing intervals. The voltage signal Vtemp can be determined from a relatively
simple integration of a time interval related to the time interval Ttemp. The voltage
signal Vprs is derived by low pass filtering the square waves of the DCAPS signal
representing the time intervals Tprs and Ttemp to obtain the average voltage.

The temperature related capacitance changes are specified to be in a narrow
range of 37 °C ±5°C which could effect an ideal gas law pressure variation of 20 mm
Hg fall scale over the 10°C temperature change. The limited range of the A/D
conversion provided by the ADC/MUX circuit 142 and the trimmed slope of the

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temperature channel integrator causes a Ttemp to range between 66 msec to 1 16 msec
in a first range and between 96.5 msec to 146.5 msec in a second range. The
resulting voltage range of the analog signal Vtemp produced at the output of the
temperature processing channel is specified to be from 0 to 1.2 volts to be processed
by the ADC/MUX circuit 142.

The blood, ambient (atmospheric, altitude, meteorologic) pressure changes
affecting the pickoff capacitor are specified in a preferable total range of 400 to 900
mm Hg. The DAC offset adjustment allows the pressure system to be adjusted under
user and/or software control to provide this total range in order to be compatible with
the more limited range of the 8 bit A/D converter.

In practice the gain of the pressure system will be adjusted dependent on the
sensitivity of the particular pressure sensor in order to provide an A/D pressure
"range" that encompasses the expected blood pressure range of the patient plus
expected local meteorologic pressure changes and expected altitude pressure changes
seen by the patient, the blood pressure is normally expected to be -10 mm Hg to 140
mm Hg of gauge pressure (relative to ambient).

The resulting voltage range of the analog signal Vprs produced at the output of
the absolute pressure signal processing channel is also specified to be from 0 to 1 .2
volts to be processed by the ADC/MUX circuit 142.

Turning again to the demodulator circuit 150 of Figure 13, it receives a
number of biasing and command signals from the digital controller/timer circuit 132,
supplies the voltages VDD and VREG to the pressure and temperature signal
modulating circuit 200, processes the sensor current pulses P R and P P , and provides
the analog signals Vtemp and Vprs to the digital controller/timer circuit 132.
Commencing first with the biasing and operating input signals, the demodulator circuit
150 receives the regulated voltage signal VREF1 at + 1.2 volts, a current signal Iin of
20 nA, and a command signal PSR ON at the power supply 156. The regulated
voltage VREF1 is the same reference voltage as is employed by the ADC/MUX
circuit 142 for digitizing the analog voltage signal between 0 and +1.2 volts into an 8-
bit digital word having 0 - 255 values. The output signals Vprs and Vtemp therefore
must fall in this range of 0 - 1 .2 volts to be processed. It is simpler then to develop
accurate regulated voltages and currents of the demodulator 150 from that same

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regulated voltage. In addition, it should be noted that the demodulator circuit 150 as
well as the other circuits of the monitor 100 including the microcomputer circuit 114
are referenced to VSS or battery ground which is at 0 volts. Therefore, the
conventions are reversed from those prevailing in the sensor circuit 200. It will be
5 understood that the same convention could be used in both cases.

From this source, the power supply 156 develops the voltage VREF2 at -2.0
volts below VDD (VDD-2.0 volts), VREG1 at +2.0 volts, and the regulated current
signals lac, Iampl, Iamp2, Iamp3, and lie that are applied to the circuit blocks of
Figure 13. The off-chip capacitor and resistor networks 155, 157 and 159, 161

10 provide bias controls ITEMP for the current lie and ISET for the current lac,
respectively. The resistor 159 is selected to provide the lac current to develop
specific current thresholds described below for the AC current sensor 154. The
resistor 157 is trimmed at the manufacture of each monitor 100 to provide a specific
current level lie for the integrator controller 174.

15 The POR and 32 kHz clock signals are applied on power-up of the monitor

100. The command signal PSR ON, and other command signals TMP ON, 2-BIT
GAIN, 4-BIT GAIN, 8-BIT DAC CNTRL, SELF CAL, PLRTY and RANGE are
provided to the demodulator circuit 150 by the digital timer/controller circuit 132 from
memory locations within the microcomputer circuit 114. Programmed-in commands

20 dictate the operating states and parameters of operation reflected by these command
signals. Command signal values and states are stored in microcomputer 114 in
memory locations that are accessed from the digital timer/controller circuit 132 and
supplied to the demodulator circuit 150 in three words. A GAIN word of 6 bits (2-
BIT GAIN & 4-BIT GAIN), a DAC word of 8 bits and a CONTROL word of 6 bits

25 (POR, PSR ON, TMP ON, SELF CAL, RANGE, PLRTY) are stored to set the
operating states and selected parameters of operation.

For example, the pressure and temperature sensing functions can be separately
programmed ON or OFF or programmed ON together by the PSR ON and TMP ON
signals. The 1-bit PSR ON command enables the bias currents Iampl, Iamp2,

30 Iamp3 to operate the pressure signal processing channel. The 1-bit TMP ON
command enables the integrator controller 174 to operate the temperature signal

WO 96/26670 29 PCT/US96/00975

processing channel. The remaining command values and states will be explained in
context of the components of the demodulator circuit 150.

Turning to the processing of the sensor current pulses P P and P R , the lead
conductor 14 is connected to the connector block terminal IS which is also connected
5 to VDD. The lead conductor 16 is connected to the VREG connector block terminal
17. A load resistor 153 is coupled across connector block terminals 15 and 17 and
between VDD and VREG in order to obtain a 2.0 volt drop and to reduce the effects
associated with changes in the lead leakage resistance 202. The lead conductor 16 at
connector block terminal 17 is connected through resistors 151 and 152 to one input

10 terminal CPIN of the AC current sensor 154 and through resistor 151 alone to the
output terminal CPOUT connected to a current sink in the AC current sensor 154.

A further input terminal of AC current sensor 154 is connected to the voltage
VREF2 at (VDD-2.0) volts developed by power supply 156. The current sensor 154
operates as a voltage regulator for ensuring that the voltage at CPOUT remains at

15 " VREF2 or (VDD-2.0) volts at all times, regardless of the effect of the current pulses
P P and P R generated during charge of the capacitors C P and C R as described above and
appearing on conductor 16 at connector block terminal 17. Since the voltage drop
across resistor 151 is small, VREG of the circuit 200 in Figure 11 may be viewed as
VREF2 of the demodulator circuit 150 of Figure 13. Resistor 151 provides protection

20 against external overdrive due to electromagnetic interference or cardioversion/
defibrillation pulses.

The current sensor 154 also includes comparators established by the current
lac that discriminate the amplitudes of current pulses P P and P R when they appear and
generate the output signals CAPSOUT and RESETOUT. The signal amplitudes are

25 discriminated and reduced to current levels established by the comparators and
reference current sources in AC current sensor 154. The CAPS_OUT signal is
developed in response to both of the low and high amplitude current pulses P R , and the
RESET_OUT signal is developed in response to the high amplitude current pulse P R
only.

30 The discrimination of the distinguishing parameters of the current pulses P P

(8.0 mA) and P R (8.0 mA followed by 24.0 mA) is effected by amplitude comparators
in AC current sensor 154 that are set by current lac provided by power supplies 156.

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The resistor 159 determines the current ISET which in turn determines the current lac
and the thresholds for the input current pulses in the AC current sensor 154.
Preferably, a low current threshold I L of +3.6 mA and a high current threshold I H of
+ 14.4 mA are established for the 8.0 mA and 24.0 mA nominal current pulse
amplitudes. The ratio of these two thresholds cannot be changed, but their values are
set by resistor 159 to allow for variances in the actual peak step amplitudes of the
current pulses P P and P R .

A sensor current pulse P P or P R having a step that exceeds the I L (+3.6 mA)
low threshold generates an output signal at CAPSOUT, whereas the high step of
current pulse P R that exceeds the I„ (+ 14.4 mA) high threshold generates an output
signal at RESET_OUT. The CAPS_OUT and RESETOUT signals are applied to the
level shifter 158 which responds by normalizing the signals between VSS or 0 volts
and VREG1 of +2.0 volts and providing the NCAPS_OUT and NRESET OUT
signals shown in Figure 12. The normalized NCAPSOUT and NRESETOUT
signals are applied to the clock and reset inverting inputs, respectively, of flip-flop
160. The inverting inputs effectively invert the depicted NCAPS OUT and
NRESET OUT signals shown in Figure 12. The flip-flop 160 responds by providing
a square wave output signal CAPS (not shown in Figure 12) at its Q-output that is
high during the interval Tprs and low during the interval Ttemp.

In the decoding of the Ttemp and Tprs intervals from the current pulse peaks
P P and P R , the NCAPS OUT signal is applied to inverting clock input of flip-flop 160
to cause it to switch state. The NRESET OUT signal is applied to the inverting reset
input of flip-flop 160 and does not cause it to change state when its state at the Q
output is already low. If, however, the Q output state is high on arrival of
NRESET_OUT, the flip-flop 160 state is switched low, resulting in the high state of
the DC APS square wave signal as shown in the first instance in Figure 12. The high
amplitude phase of current pulse P R therefor synchronizes the state of the DCAPS
square wave signal on power up and restores any loss of synchronization that may
occur from time to time. Once synchronization is established, each successive 8 mA
step of the respective current pulse peaks P P and P R shown in Figure 12 switches the
Q output state of the flip-flop 160, causing the square wave of the CAPS and DCAPS
signals reflecting the Ttemp and Tprs intervals.

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The CAPS square wave output signal is applied to a digital signal processor
162 and is normally inverted to provide the DCAPS signal shown in Figure 12 at a
first output. The digital signal processor 162 also normally inverts the CAPS signal to
provide the TREF signal at a second output. In this fashion, the Tprs interval of
5 DCAPS provided to the input of pressure signal processing channel 163 is negative in
polarity, and the Ttemp interval of TREF provided to the input of the temperature
signal processing channel 164 is positive in polarity. In regard to the polarities of
signals DCAP and TREF, the digital signal processor 162 also receives the PLRTY
signal from the digital controller/timer circuit 132. The PLRTY signal may be

10 selectively programmed to invert the polarity of the DCAPS square wave in order to
increase the operating range of the pressure signal processing channel 163. However,
it is expected that the PLRTY signal would seldom be changed, and such a
programming option may be eliminated if the range provided in the pressure signal
processing channel 163 is sufficient.

15 As described further below, a self calibration mode can be initiated in response

to a SELF CAL signal to apply a 5.46 kHz square wave signal through the digital
signal processing circuit 162 to the temperature integrator controller 174 for
calibration purposes. The 5.46 kHz square wave signal is simply chosen for
convenience, since it is an even sub-multiple of the 32 kHz clock frequency and is

20 close to the nominal 5 kHz operating frequency. The following discussion assumes
first that the temperature processing channel 164 is already calibrated in the manner
described below and that the normal operating mode is programmed (SELF CAL off)
so that only the CAPS signal is processed by the digital signal processor 162.
Addressing the derivation of the signal Vtemp by the temperature signal

25 processing channel 164 first, the temperature is demodulated from the high state of the
TREF square wave signal having a duration directly relating to the charge time
Ttemp. The integrator controller 174 employs the current lie to charge an integrator
capacitor 187 over the time Ttemp (or a portion of that time as explained below) and
then charges sample and hold capacitor 190 to the voltage on integrator capacitor 187.

3 0 The voltage on integrator capacitor 187 is then discharged and the voltage on sample
and hold capacitor 190 is amplified by temperature amplifier stage 195 to become the
Vtemp signal in the range of 0 - 1.2 volts.

WO 96/26670 32 PCT/US96/00975

More particularly, when the integrator capacitor 187 is not being charged or
the voltage transferred to the sample and hold capacitor 190, both plates of the
integrator capacitor 187 are held at VREG1 and the bidirectional switch 176 is open.
Again, the discharge state is characterized as a state where there is no net voltage or
5 charge on the capacitor 187, and the charged state is characterized by a net voltage
difference across its plates, even though the "charged" voltage may be nominally
lower than the "discharged" voltage.

When the TREF signal goes high (and the low range is programmed), the
integrator controller 174 commences charging the plate of integrator capacitor 187

10 connected to resistor 188 to a voltage lower than VREG1 through a current sink to
VSS internal to integrator controller 174. At the end of the high state of the TREF
signal, the current sink to VSS is opened and the bidirectional switch 176 is closed for
one clock cycle time (30.5 msec) to transfer the resulting voltage level on capacitor
187 to capacitor 190. Bidirectional switch 176 is then opened, and capacitor 187 is

15 discharged by setting both plates to VREG1 through switches internal to integrator
controller 174. With each successive recharge of integrator capacitor 187, the
capacitor 187 voltage level achieved varies upward and downward from its preceding
voltage level with changes in the width of the high state of the TREF signal, and the
new voltage level is transferred to capacitor 190. The new voltage level is held on

20 capacitor 190 when the switch 176 is opened.

The switching of bidirectional switch 176, resistor 188 and capacitor 190 also
form a low pass filter. The pass band of this filter is sufficient to allow only the
temperature related component of the signal to pass through and be reflected on
capacitor 190.

25 The resulting voltage on capacitor 190, amplified by amplifier stage 195,

provides the Ytemp signal representing the temperature in the pressure sensor cavity.
Amplifier stage 195 includes an amplifier 178 referenced back to approximately
+ 1.2V through the voltage divider comprising resistors 191, 192, 193 dividing the
VREG1 of +2.0 volts. Amplifier stage 195 has a gain of two, and so the maximum

30 voltage which the sample and hold capacitor 190 can reach is +0.6V. This

corresponds to a +0.6 volt level on integrating capacitor 187 at its junction with
resistor 188 which is achieved in 116 msec employing the regulated current lie.

WO 96/26670 33 PCT/US96/00975

Two operating ranges provide a higher resolution of the possible values of the
reference capacitor R charging time Ttemp reflected by the high state of the TREF
signal. Either a high or low range must be programmed by the RANGE bit based on
individual sensor circuit 200 characteristics and/or the temperature range of the
5 patient. Since a 5° C change in temperature will result in approximately 5% change
in Ttemp, the 8-bit ADC count provided by ADC/MUX circuit 142 in response to
Vtemp for a particular lead cannot be near the limits of 0 and 255.

For this reason, both the high range and low range for the temperature are
provided, and one or the other is selected via a one-bit value of the above-referenced

10 6-bit CONTROL word. Setting the RANGE bit to 1 places integrator controller 174
in the high range mode which corresponds to a TREF high state pulse width of 96-146
msec. Programming the RANGE bit to 0 places integrator controller 174 in the low
range mode which corresponds to a TREF high state pulse width of 66-116 msec. The
limit of 0.6 volts can be reached at the upper end of this pulse width range.

15 However, it is anticipated that the high operating range will be necessary in

certain instances. When the high range mode is selected, the integrator controller 174
effectively prolongs the high state TREF square wave by delaying the charging of the
integrator capacitor 187 by one clock cycle or 30.5 msec from the beginning of the
high state TREF square wave. This effectively shortens the TREF high state pulse

20 width range of 96-146 msec that is integrated back to 66-116 msec, allowing the
Vtemp voltage signal to fall into the 0 - 0.6 volt range that can be doubled in
amplifier stage 195, digitized and stored. The programmed range is also stored with
the digitized temperature data so that the proper values can be decoded from the
telemetered out data.

25 In order to set the RANGE for proper temperature measurement in a given

patient, one or the other range is programmed and the digitized temperature readings
are accumulated and telemetered out. If they are in a proper range, then the
programmed RANGE is correct. In general, if in the low range mode and if the
digital temperature value is a digital word 50 or less, it is necessary to program the

3 0 high range. And, if in high range mode and the digital word is 200 or more, it is
necessary to program to the low range. Alternatively, the range could be
automatically switched at these threshold levels.

WO 96/26670 34 PCT/US96/00975

The rate of charge of integrator capacitor 187 in these ranges to get to the
proper voltage range of 0 - 0.6 volts depends on the current lie. The self calibration
of the temperature signal processing channel 164 is necessary to trim the resistor 157
to precisely set the current ITEMP and the current lie so that a voltage of 0.590 volts
5 is reached on capacitor 187 after a 116 msec integration time. In this mode, the

RANGE is programmed to the low range, and the SELF CAL signal is programmed
ON. The digital signal processor 162 responds to the SELF CAL ON signal to divide
the 32 kHz clock signal provided from digital controller/timer circuit 132 by 6 into a
5.46 kHz square wave signal exhibiting a 50% duty cycle. The digital signal

10 processor substitutes the square wave calibration signal for the TREF signal and
applies it to the temperature signal processing channel 164 and to the input of the
integrator controller 174. The resistance of resistor 157 is trimmed to adjust
integrator current lie until the voltage 0.590 volts is achieved in 116 msec or an ADC
count of 125 is reached.

15 Turning now to the derivation of the pressure signal Vprs, the nominally 5

kHz DCAPS positive and negative square wave of + 2.0 volts is filtered and averaged
to derive a voltage signal Vprs in the range of 0 - 1.2 volts at the junction of capacitor
182 and resistor 186. The 5 kHz signal component is filtered out by a 4-pole filter
including a 250 Hz low pass filter provided by capacitor 165 and resistor 166, an

2 0 active Butterworth filter comprising 40 Hz low pass filter network 196 and first
pressure amplifier stage 197, and a further 1 pole, 250 Hz low pass filter pole
comprising capacitor 182 and resistor 186. The low pass filter network 196 comprises
the resistors and capacitors 165 -167, 169, 171, 173, 175, 182 and 186 and averages
the voltage square wave to create a D.C. voltage proportional to the DCAPS square

25 wave signal duty cycle. The first pressure amplifier stage 197 buffers the filtered
pressure-related signal at its output. The filtered output signal is applied to second,
inverting, pressure amplifier stage 198 which comprises the amplifier 170 and the
programmable gain, switched resistor networks 180 and 181. Amplification and
voltage offset of the output signal of amplifier 168 is provided in second pressure

30 amplifier stage 198 by the 2-BIT GAIN, 4-BIT GAIN and 8-bit offset DAC settings.

The variations in the manufacturing tolerances and conditions of the sensor
module 20 affects the reference and pickoff capacitance values and the response to

WO 96/26670 35 PCT/US96/00975

temperature and pressure changes that particularly affect the pressure sensing function.
The gain and offset adjustments are provided to correct for such affects. The offset
adjustment is also required to provide for pressure range adjustments so that the
pressure range used provides adequate resolution of pressure differences. As
5 mentioned above, the A/D conversion range of the ADC/MUX circuit 142 is limited
to 256 digitized values from a voltage range of 1.2 volts. Therefore, it is necessary to
compensate for variations in the patient's own blood pressure range as well as
prevailing atmospheric pressure primarily related to the altitude that the patient
normally is present in. These compensations are included in an offset factor

1 0 developed at the time of implant.

The offset factor is provided by the 8-bit offset digital to analog converter
(DAC) 172 which provides an offset analog voltage dependent on the programmed
value of the DAC CNTRL binary coded digital word. The primary function of the
DAC 172 is to provide the analog voltage to H zero" the offset in the system in order to

15 keep the pressure signal within the range of the ADC/MUX block 142 (0-1 .2 volts).
The analog offset voltage is applied to differential pressure amplifier 170 where it is
subtracted from the output voltage of the first pressure amplifier 168. The total
programmable range of the DAC 172 is 630 mV, between 570 mV to 1,200 mV.

The gain settings for the second pressure amplifier stage 198 can be adjusted

20 by programming values for the 4-BIT GAIN and 2-BIT GAIN binary words stored in
the RAM 124 of Figure 1 by the external programmer. The 4-BIT GAIN signal
controls the gain of the pressure amplifier stage 198 by setting switched resistors in
feedback switched resistor network 180 to the binary coded gain word to provide a
gain range that is selectable by the further 2-BIT GAIN signal setting of switched

2 5 resistor network 181 . The gain setting can be varied from 5X - 20X in IX

increments, 10X - 40X in 2X increments and 20X - 80X in 4X increments. The gain
ranges and increments are established by the 2-BIT GAIN control signal applied to the
series switched resistor network 181.

The first pressure amplifier stage 197 responds to the ratio of Ttemp to the

3 0 sum of Ttemp and Tprs resulting in a first filtered voltage signal. The second

pressure amplifier stage 198 amplifies and inverts the first voltage signal as a function
of the offset and gain settings and therefore responds effectively to the ratio of Tprs to

WO 96/26670 36 PCT/US96/00975

the sum of Ttemp and Tprs, or the duty cycle of Tprs. The output signal from
amplifier 170 of the second amplifier stage 198 is applied to a further low pass filter
stage comprising resistor 186 and capacitor 182 to filter out any remaining component
of the about S kHz oscillation frequency and/or any noise. The resulting filtered
5 signal is applied as pressure signal Vprs to the digital controller/timer circuit 132 of
Figure 1 .

The resulting Vprs and Vtemp voltage signals are digitized in the ADC/MUX
circuit 142 in a manner well known in the art to provide digitized Vprs and Vtemp
data values. The digitized Vprs and Vtemp data values are applied on bus 130 to the

10 microcomputer circuit 114 for storage in specified registers in RAM/ROM unit 128.
The digitized Vtemp data value may be employed in processing the digitized data
values telemetered out by the external programmer to compensate for the temperature
induced affects on the Vprs data values. The stored data values may also be
correlated to data from the activity sensor block 152 and other sensors, including other

15 lead borne sensors for monitoring blood gases and the patient's EGM as described
above.

The capacitive pressure and temperature sensing system described above is
intended for implantation in the body of a patient. However, it will be understood that
the sensor lead 12 may be implanted as described above through a venous approach

2 0 but with its proximal connector end coupled through the skin to an external system
100 for ambulatory or bedside use. In addition, the system 100 may be simplified to
the extent that the data may be transmitted remotely in real time to an external
programmer/transceiver instead of being stored in microcomputer circuit 114. In the
latter case, the microcomputer circuit 1 14 may be eliminated in favor of a more

25 limited, digital discrete logic, programming command memory of types well known in
the prior art of pacing.

Variations and modifications to the present invention may be possible given the
above disclosure. Although the present invention is described in conjunction with a
microprocessor-based architecture, it will be understood that it could be implemented

30 in other technology such as digital logic-based, custom integrated circuit (IC)
architecture, if desired.

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It will also be understood that the present invention may be implemented in
dual-chamber pacemakers, cardioverters, defibrillators and the like. However, all
such variations and modifications are intended to be within the scope of the invention
claimed by this letters patent.

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CLAIMS

1 An implantable capacitive pressure sensor lead for providing signals
representative of the magnitude of body fluid pressure at a selected site and ambient
operating conditions, including body temperature, at the site comprising:

an elongated implantable lead body having proximal and distal end sections and
having first and second electrical conductors extending from the proximal end section
to the distal end section, said proximal end adapted to be coupled to a biasing signal
source, and said distal end section adapted to be implanted in a body position for
measuring a varying body fluid pressure; and

a pressure sensor module formed in the distal end section of the lead body and
coupled to said first and second electrical conductors, the pressure sensor module

further comprising:

a module housing attached to said lead body adapted to be positioned in
a body cavity and enclosing a hermetically sealed chamber;

pressure deformable pickoff capacitor means for varying in capacitance
in response to variations in fluid pressure having a first plate formed of a
pressure deformable diaphragm of said module housing and a second plate
spaced apart therefrom;

reference capacitor means having a first plate formed of a non-
deformable portion of said module housing and a second plate spaced apart
therefrom;

a substrate supported within said module housing for supporting said

second plates; and

pressure and temperature signal modulating circuit means supported on

said substrate within said hermetically sealed chamber and electrically coupled

to said first and second electrical conductors.

2. The pressure sensor lead of Claim 1 wherein:

said first plate of said pickoff capacitor means is formed of a pressure
deformable, planar diaphragm having a first predetermined surface area formed by
said module housing of a conductive material with an exterior planar surface and a
parallel interior surface within said hermetically sealed chamber, said planar

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PCT/US96/00975

diaphragm adapted to be deformed in said first predetermined surface area by
variations in fluid pressure outside the module housing; and

said first plate of said reference capacitor means is formed by a second
predetermined planar surface area of said interior surface spaced apart from and co-
planar with said first predetermined surface area and substantially non-deformable by
variations in fluid pressure outside the module housing.

3. The pressure sensor lead of Claim 2 Anther comprising:

means for forming said second plates of said pickoff and reference capacitor
means on a first surface of said substrate; and

means for spacing said second plates of said pickoff and reference capacitor
means a fixed distance from said first plates of said pickoff and reference capacitor
means.

4. The pressure sensor lead of Claim 3 wherein:

said second plates of said pickoff and reference capacitor means further
comprise pickoff and reference capacitor platings of a first thickness formed on said
first surface of said substrate conforming to said pickoff and reference capacitor plate
patterns, respectively; and
said spacing means further comprises:

a plurality of standoffs of a second thickness greater than said first thickness
formed on said first surface of said substrate; and

means for urging said standoffs against said interior surface for spacing said
second plates of said pickoff and reference capacitor means from said first plates of
said pickoff and reference capacitor means by a predetermined gap establishing the
same capacitance for both the pickoff and reference capacitors in the absence of
deformation of said diaphragm.

5. The pressure sensor lead of Claim 3 wherein said standoffs are formed
of standoff platings of said second thickness formed on said first surface of said
substrate in areas outside said pickoff and reference capacitor platings and electrically
isolated therefrom.

6. The pressure sensor lead of Claim 4 wherein said pressure and
temperature signal modulating circuit means is supported on a second surface of said

WO 96/26670 40 PCT/US96/00975

substrate within said hermetically sealed chamber and electrically coupled to said first
and second electrical conductors and to said pickoff and reference capacitor platings.
7. The pressure sensor lead of Claim 1 further comprising:
means for supporting said second plates of said pickoff and reference capacitor
5 means on a first surface of said substrate; and

means for spacing said second plates of said pickoff and reference capacitor
means a fixed distance from said first plates of said pickoff and reference capacitor
means whereby a predetermined gap establishes the same capacitance for both the
pickoff and reference capacitors in the absence of deformation of said diaphragm.
10 8. The pressure sensor lead of Claim 7 wherein said pressure and

temperature signal modulating circuit means is supported on a second surface of said
substrate within said hermetically sealed chamber and electrically coupled to said first
and second electrical conductors and to said second plates.

9. The pressure sensor lead of Claim 1 wherein:

15 said module housing is formed of a conductive material having an exterior

planar surface and an interior planar surface shaped to form said planar diaphragm
having a fixed thickness and perimeter boundary in a planar section thereof to form
said first deformable plate of said pickoff capacitor in a pickoff capacitor plate pattern,
said interior planar surface extending beyond the boundary of said planar diaphragm to

20 form said first plate of said reference capacitor in a reference capacitor plate pattern
adjacent to said first deformable plate.

10. The pressure sensor lead of Claim 9 further comprising:

means for forming said second plate of said pickoff capacitor means on a first
surface of said substrate, said second plate of said pickoff capacitor means shaped to

2 5 correspond to said pickoff capacitor plate pattern;

means for forming said second plate of said reference capacitor means on a
first surface of said substrate, said second plate of said reference capacitor shaped to
correspond to said reference capacitor plate pattern; and

means for spacing said second plates of said pickoff and reference capacitor

3 0 means a fixed distance from said first plates of said pickoff and reference capacitor

means.

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1 1 . The pressure sensor lead of Claim 10 wherein:

said second plates of said pickoff and reference capacitor means further
comprise pickoff and reference capacitor platings of a first thickness formed on said
first surface of said substrate conforming to said pickoff and reference capacitor plate
5 patterns, respectively; and

said spacing means further comprises:

a plurality of standoffs of a second thickness greater than said first thickness
formed on said first surface of said substrate; and

means for urging said standoffs against said interior surface for spacing said
10 second plates of said pickoff and reference capacitor means from said first plates of
said pickoff and reference capacitor means by a predetermined gap establishing the
same capacitance for both the pickoff and reference capacitors

12. The pressure sensor lead of Claim 1 1 wherein said standoffs are formed
of standoff platings of said second thickness formed on said first surface of said

1 5 substrate in areas outside said pickoff and reference capacitor platings and electrically
isolated therefrom.

13. The pressure sensor lead of Claim 10 wherein said pressure and
temperature signal modulating circuit means is supported on a second surface of said
substrate within said hermetically sealed chamber and electrically coupled to said first

20 and second electrical conductors and to said pickoff and reference capacitor platings.

14. The pressure sensor lead of Claim 1 wherein said module housing is
fabricated of a conductive metal and further comprising:

insulative body compatible sheath means for electrically isolating said module
housing from body fluids and tissue.
25 15. The pressure sensor lead of Claim 14 wherein said sheath means further

comprises an insulative, uniform thickness coating of an adhering body compatible
material overlying said diaphragm.

16. The pressure sensor lead of Claim 14 wherein said sheath means further
comprises a tubular sheath fitting over said module housing and coupled to said
30 insulative, uniform thickness coating of an adhering body compatible material
overlying said diaphragm.

WO 96/26670

42

PCT/US96/00975

17. The pressure sensor lead of Claim 1 wherein said pressure and
temperature signal modulating circuit means further comprises:

means for receiving an externally applied biasing signal on said first and
second conductors and for generating charging and discharging currents therefrom
which vary in magnitude with ambient temperature;

means for alternately charging and discharging said pickoff and reference
capacitor means with said charging and discharging currents; and

timing pulse generating means for generating pickoff and reference timing
pulses on one of said electrical conductors, the timing pulses separating the alternate
charging time intervals of the pickoff and reference capacitor means.

18. The pressure sensor lead of Claim 17 wherein said timing pulse
generating means further comprises:

means for establishing a charge threshold voltage;

means for detecting the charging of said reference capacitor means to said
charge threshold voltage and for generating a pickoff timing pulse on said first
electrical conductor in response thereto, said pickoff timing pulse having a first
predetermined characteristic;

means for detecting the charging of said pickoff capacitor means to said charge
threshold voltage and for generating a reference timing pulse in response thereto, said
reference timing pulse having a second predetermined characteristic distinguishable
from said first predetermined characteristic.

19. An implantable capacitive pressure sensor lead for providing signals
representative of the magnitude of body fluid pressure at a selected site and ambient
operating conditions, including body temperature, at the site comprising:

an elongated implantable lead body having proximal and distal end sections and
having first and second electrical conductors extending from the proximal end section
to the distal end section, said proximal end adapted to be coupled to a biasing signal
source, and said distal end section adapted to be implanted in a body position for
measuring a varying body fluid pressure; and a pressure sensor module formed in
the distal end section of the lead body and coupled to said first and second electrical
conductors, the pressure sensor module further comprising:

WO 96/26670 43 PCT/US96/00975

a module housing attached to said lead body adapted to be positioned in
a body cavity and enclosing a hermetically sealed chamber;

pressure deformable pickoff capacitor means for varying in capacitance
in response to variations in fluid pressure having a first plate formed of a
5 pressure deformable diaphragm of said module housing and a second plate

spaced apart therefrom;

reference capacitor means having a first plate formed of a non-
deformable portion of said module housing and a second plate spaced apart
therefrom; and

10 pressure and temperature signal modulating circuit means within said

hermetically sealed chamber and electrically coupled to said first and second
electrical conductors.

20. The pressure sensor lead of Claim 19 wherein said pressure and
temperature signal modulating circuit means further comprises:

15 means for receiving an externally applied biasing signal on said first and

second conductors and for generating charging and discharging currents therefrom
which vary in magnitude with ambient temperature;

means for alternately charging and discharging said pickoff and reference
capacitor means with said charging and discharging currents; and

20 timing pulse generating means for generating pickoff and reference timing

pulses on one of said electrical conductors, the timing pulses separating the alternate
charging time intervals of the pickoff and reference capacitor means.

21 . The pressure sensor lead of Claim 20 wherein said timing pulse
generating means further comprises:

25 means for establishing a charge threshold voltage;

means for detecting the charging of said reference capacitor means to said

charge threshold voltage and for generating a pickoff timing pulse on said first

electrical conductor in response thereto, said pickoff timing pulse having a first

predetermined characteristic;
3 0 means for detecting the charging of said pickoff capacitor means to said charge

threshold voltage and for generating a reference timing pulse in response thereto, said

WO 96/26670

44

PCT/US96/00975

reference timing pulse having a second predetermined characteristic distinguishable
from said first predetermined characteristic.

22. The pressure and temperature sensing lead of Claim 21 wherein:
said first predetermined characteristic comprises a first pulse width, pulse

amplitude and pulse leading edge slope; and

said second predetermined characteristic comprises a stepped pulse having
leading and trailing steps, said leading step corresponding to said first pulse width,
pulse amplitude an pulse leading edge slope, and said trailing step having a second
pulse amplitude and pulse width.

23. A method of fabricating an implantable capacitive pressure sensor lead
for providing signals representative of the magnitude of body fluid pressure at a
selected site and ambient operating conditions, including body temperature, at the site
comprising:

forming an elongated implantable lead body having proximal and distal end
sections with first and second electrical conductors extending from the proximal end
section to the distal end section, said proximal end adapted to be coupled to a biasing
signal source, and said distal end section adapted to be implanted in a body position
for measuring a varying body fluid pressure;

covering said first and second electrical conductors with a first body
compatible, electrically insulating outer sheath extending from said proximal end
section to said distal end section;

attaching a pressure sensor module to the distal ends of said first and second
electrical conductors at said distal end section, the pressure sensor module further
comprising:

a module housing enclosing a hermetically sealed chamber formed of a
conductive material;

pressure deformable pickoff capacitor means for varying in capacitance
in response to variations in fluid pressure having a first plate formed of a
pressure deformable diaphragm of said module housing and a second plate
spaced apart therefrom; and

WO 96/26670

45

PCT/US96/00975

pressure and temperature signal modulating circuit means within said
hermetically sealed chamber and electrically coupled to said first and second
electrical conductors;

covering said sensor module with a second sheath of the same material as said
5 first sheath, the material of said second sheath having a low adherence to the material
of said module housing;

providing an opening in said second sheath corresponding to the area of said
diaphragm;

bonding said second sheath to said first sheath in a fluid tight bond at the
0 junction of said distal end of said lead body with said module housing so that said
sheath opening is aligned with said diaphragm;

filling the gap between the perimeter of the opening of the second sheath and
the module housing with said further material to decrease the migration of body fluids
into the space between the module housing and the second sheath; and
5 applying a further body compatible, electrically insulating material in a

uniform thickness layer over said diaphragm in said opening of said second sheath,
said further material having a high adherence with the material of said diaphragm in
the presence of body fluids and tissue.

24. The method of Claim 23 wherein said material of said first and second
0 sheathes is an implantable grade polyurethane and the further body compatible
material is an implantable grade silicon rubber.

WO 96/26670

PCT/US96/00975

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INTERNATIONAL SEARCH REPORT

I Men* oruJ Application No

PC i /US 96/00975

A. CLASSIFICATION OF SUBJECT MATTER

IPC 6 A61B5/O0

According to International Patent Clasaficaoon (IPC) or to both national classification and IPC

B. FIELDS SEARCHED

Minimum documentation searched (d&tsihcabon system followed by classification symbols)

IPC 6 A61B

Documentation searched other than minimum documentation to the extent that such documents art inducted in the fields searched

Electronic data base consulted during the international search (name of data base and* where practical, search terms used)

C. DOCUMENTS CONSIDERED TO BE RELEVANT

Category *

Citation of document, with indication, where appropriate, of the relevant passages

Relevant to claim No.

x
x

US.A.4 815 472 (K.D. WISE ET AL.) 28 March
1989

see column 2, line 58 - column 4, line 20
see column 5, line 1 - column 7, line 22
see column 9, line 21 - column 10, line 14
see column 12, line 39 - column 14, line
51

-/--

1-3,6,8

9,13-15
17-21,23

Further documents are listed tn the continuation of box C.

0

Patent family me m ber s are listed in annex.

* Special categories of ated documents :

"A" document defining (he general state of the art which is not
considered to be of particular relevance

"E* earlier document but published on or after the international
filing date

*L* document which may throw doubts on pnonty daim(s) or
wtech ts ated to establish the publication date of another
citation or other special reason (as xpcciQed)

"O* document referring to an oral disclosure, use, exhibition or
other means

*P' document published pnor to the intemaoonal filing date but
later than the pnonty date claimed

T later document published after the international filing dale
or pnonty date and not in conflict with the application but
died to understand the pnnctplc or theory underlying the
invention

'X* document of particular relevance; the claimed invention
cannot be considered novel or cannot be considered to
involve an inventive step when the document is taken alone

* Y* document of particular relevance; the claimed invenbon
cannot be considered to involve an inventive step when the
document ts combined with one or more other such docu-
ments, such combination being obvious to a person skilled
tn the art.

'&* document member of the same patent family

Date of the actual completion of the intemaoonal search

16 July 1996

Date of mailing of the international search report

21.08.96

Name and mailing address of the ISA

European Patent Office, P.B. S&IS Patcntlaan 2
NL - 22S0 HV Ripwtjk
Td. ( - 31-70) 340-2040, Tx. 31 651 epo nL
Fax (* 31-70) 340-3016

Authorized officer

Rieb, K.D.

Form PCT.1SA,-2IO(i

page 1 of 2

INTERNATIONAL SEARCH REPORT

CiCononuaflon) DOCUMENTS CONSIDERED TO BE RELEVANT

Category '

CiUtioo of oocumcnu with indication, where Appropriate, ol the relevant passagci

SENSORS AND ACTUATORS,
vol. A41, no. 1/3, April 1994, LAUSANNE
(CH),

pages 198-206, XPO0O450031
P. WOUTERS ET AL.: "A low power
multi -sensor interface for injectable
microprocessor-based animal monitoring

system"

see page 198, right-hand column, paragraph
2 - page 199, left-hand column, paragraph

see page 203, right-hand column, paragraph
2 - page 205, right-hand column, paragraph
2

US,A,4 237 900 (J.H. SCHULMAN ET AL.) 9
December 1980

see column 3, line 52 - column 4, line 38
see column 7, line 60 - column 8, line 57

US,A,5 067 491 (M.S. TAYLOR, II. ET AL.)
26 November 1991
see column 2, line 28 - line 51
see column 3, line 53 - column 4, line 59

Intcrr ma) Application No

PCI /US 96/00975

Relevant to clam No,

1-3,19

1-3,19

1,19,24

F«> rcrmA.ua ******** <* n«t) u«iy im)

page 2 of 2

INTERNATIONAL SEARCH REPORT

.nformauon on patent family members

Intr ona) Application So

PC 1 /US 96/00975

Patent document
cited in search report

Publication
date

Patent family
member (i)

Publication
date

US-A-4815472 28-03-89 US-A- 4881410 21-11-89

US-A- 5013396 07-05-91

US-A- 5113868 19-05-92

US-A- 5207103 04-05-93

US-A-4237900

09-12-80

NONE

US-A-5067491

26-11-91

NONE

Form PCT/1SA/2J0 (pttaM family mm) (July 1993)

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